Interstage coupling networks couple signal from the output of one stage to the input of another stage and usually include frequency selective tunable circuits for selecting the pass band of the coupled signal. Interstage coupling networks are typically included in the tuner portion of a television receiver for selecting the signals which are passed from an RF signal amplifying stage to a heterodyning mixer stage. The interstage coupling network includes tunable elements which are adjusted in response to the selected television channel for passing the proper frequency range of the received RF signals to the mixer so that the selected television signals can be processed for ultimately presenting a video and audio program to a viewer. In general, television receivers must be able to select a particular RF signal for a specified channel out of several different frequency ranges, or bands, of broadcast television signals. By way of example, in the United States, the television tuner must be capable of selecting channels for which the RF picture carrier frequency may be in the low-VHF, high-VHF or the UHF tuning band.
Continuous tuning of an interstage coupling network from the low-VHF through to the UHF frequency band is generally not possible of the great frequency range which must be covered. Therefore, television tuner interstage coupling networks typically include switching provisions for selecting different reactive elements for inclusion therein, depending on the frequency band of the selected channel. FIG. 1a illustrates a prior art tunable interstage coupling network suitable for use in a multiband television tuner. A tunable interstage coupling network 10 couples RF signals from a signal source 12 to a load circuit 14. Source 12 may comprise the output of an FET RF amplifying stage, represented by a current source i.sub.s coupled in parallel with an impedance R.sub.p, which receives television RF signals from an antenna (not shown) and supplies the full range of received RF signals to an input terminal 16 of network 10. Load circuit 14 may comprise the input impedance of a heterodyning mixer circuit and is represented by a resistor R.sub.L which receives the selected RF signals from output terminal 18 of network 10. Network 10 includes a primary inductor 20 having one end coupled to input terminal 16 and a secondary inductor 22 having one end coupled to output terminal 18. The other ends of inductors 20 and 22, respectively, are coupled to the anodes of series-connected, oppositely-poled diodes 24 and 26. The junction of diodes 24 and 26 is coupled to a point of reference potential, such as signal ground, via a series connection of a mutual inductor 28 and a first source of switching signal V.sub.s1. A series connection of two further inductors 30 and 32 is coupled in parallel with series-connected diodes 24 and 26. The junction between inductors 30 and 32 is also coupled to signal ground, via a mutual inductor 34 and a second source of switching signal V.sub.s2. Variable capacitance tuning elements 36 and 38 are coupled to terminals 16 and 18, respectively, and concurrently controlled (as indicated by the dashed lines) for tuning the RF signal passband of coupling network 10 within the frequency band selected by the polarity of switching signals V.sub.s1 and V.sub.s2.
For receiving RF signals in a first tuning band, switching signal source V.sub.s1 provides a high positive voltage to the cathodes of diodes 24 and 26 while source V.sub.s2 provides signal ground to their anodes. This biases diodes 24 and 26 into conduction and results in a configuration for interstage coupling network 10 as illustrated in FIG. 1(b). In that figure, the forward resistance of the diodes and the equivalent inductance of their signal leads are small enough to be insignificant and therefore are not shown. A coupling network of this type is generally said to have "low-side L" inductive mutual coupling, due to the fact that the coupling between inductors 20 and 22 is substantially determined by an inductor 28, which is coupled from the junction of inductors 20 and 22 to the low side of network 10, i.e., signal ground. As is readily apparent, as the frequency of interest is increased, the impedance of inductor 28 increases, allowing for a greater coupling of signals from the primary side (input) to the secondary side (output) of network 10.
Tracking tunable elements 36 and 38 can tune a passband of e.g., 10 MHz, from the low end to the high end of the selected frequency band. The desired tuning is shown in FIG. 2 for passbands tuned at the low (f.sub.1), middle (f.sub.2) and high (f.sub.3) portions of the selected frequency band. Note that it is desirable that the bandwidth of the RF signal passband be relatively constant across the selected frequency band and that frequencies above the f.sub.3 passband be sharply attenuated. In a television receiver, frequencies just above the selected tuning band may include image frequencies, which, if not substantially attenuated, can undesirably effect the quality of the reproduced picture and/or sound. Coupling network 10 exhibits a passband bandwidth variation which varies across the selected band in accordance with the cube of the frequency (f.sup.3). It is herein recognized as desirable to compensate for this bandwidth variation by using what is generally referred to as "top-side C" capacitive coupling which comprises adding a coupling capacitor C.sub.c such as shown in FIG. 1(b), between terminals 16 and 18. As generally known to those skilled in the design of interstage coupling networks, capacitor C.sub.c reduces passband bandwidth variations to approximately 2:1 across the selected frequency band and additionally forms a resonant trap circuit with the "low-side L" inductor 28 of network 10 for attenuating the image frequency signals which are just above the upper end of the selected frequency band. This effect can intuitively be understood once it is realized that low side L coupling has a positive coefficient of coupling and top side C coupling has a negative coefficient of coupling. Therefore the top side C coupling tends to cancel the low side inductive coupling, thereby substantially compensating for the passband bandwidth variations caused by the low side inductive coupling and forms a trap circuit at the frequency wherein their coupling is equal but opposite. For the sake of simplicity, the resultant 2:1 passband variation is not shown in FIG. 2, however, the effect of the image trap is clearly indicated for signal frequencies equal to or greater than the upper end of the f.sub.3 passband.
For receiving RF signals in a second tuning band of, e.g., a lower frequency range than the first tuning band, switch signal source V.sub.s1 provides a low voltage to the anodes of diodes 24 and 26 while switch source V.sub.s2 provides a high positive potential to their cathodes. This biases diodes 24 and 26 out of conduction and results in a configuration for network 10 as illustrated in Figure 1(c). Thus, inductors 30 and 32 become part of the series connection of inductors between input and output terminals 16 and 18. Low side L inductive coupling is provided in this configuration by an inductor 34 coupled from the junction of inductors 30 and 32 to signal ground via the second source of switching signal V.sub.s2. It is herein recognized as desirable to also include top side C coupling in this configuration of network 10. Due to the lower frequency of the second tuning band, the capacitance value of a top-side C coupling capacitor C.sub.c ' must be greater than the capacitance value of C.sub.c of FIG. 1(b), in order to provide an image trap for frequencies just above the high end of the second tuning band. If C.sub.c were used in the FIG. 1(c) configuration, its capacitance value would be too low for the lower frequency tuning band of the second configuration, and the image trap would undesirably occur at a higher frequency within the tuning range of the second tuning band.
In order to provide both C.sub.c and C.sub.c ' in coupling network 10 of FIG. 1(a), it would seem appropriate to simply include them in a passband shaping network connected between terminals 16 and 18 which includes a series connection of a second capacitor and a switching diode in parallel with C.sub.c such that when the diode is conductive, the parallel combination of the second capacitor with C.sub.c would equal C.sub.c '. However, it is undesirable to include a switching diode in the top-side C coupling circuit of FIG. 1(a) due to its increased cost and added circuit complexity. Perhaps more importantly, the signal frequency of the first and second bands may be relatively high and the parasitic reactance of the switching diode may adversely effect the equivalent top-side C coupling circuit and a proper value for C.sub.c ' may not be attainable. Thus, it would be desirable to connect a coupling capacitor to network 10 such that the passband bandwidth variation is minimized and the location of the image frequency trap remains at a useful position just above the upper end of each selectable frequency band.